1. Field of the Invention
The present invention generally relates to the field of mixers. More specifically, the present invention relates to differential mixers.
2. Discussion of the Related Art
Such mixers are used in many applications, for example, transmission/reception chains of portable phones. In the present description, chains capable of being usable in a multimode fashion, that is, capable of receiving signals exhibiting different frequency characteristics, will be considered. Thus, a chain in the sense of the present invention will be capable of receiving at least two different signals chosen from among signals of UMTS or WCDMA standard of a frequency on the order of 2.16 Hz, of GSM standard of a frequency on the order of 900 MHz, or of DCS standard of a frequency of 1.8 GHz.
FIG. 1 schematically illustrates the functional structure of a portion of a receive chain.
The chain includes an input block 1 (RF LNA) for receiving a radiofrequency signal and including as many low-noise amplifiers 2 as there are possible modes of the input signal, three in the considered case. Each low-noise amplifier 2 is associated with a selection switch 3. The control of switches 3 is performed by a common control circuit (not shown). The amplified radiofrequency signal RF coming out of block 1 is mixed by a mixer 4 with a local oscillator signal LO to provide a signal of intermediate frequency IF. The structure of mixer 4 will be described in detail hereinafter. Then, the signal of intermediate frequency IF is filtered by a block 5 (IF FILTER). In a multimode system, several parallel filtering branches are provided, each corresponding to at least one of the possible modes. In the considered case, it is possible to only use two filtering branches, a first branch performing a filtering at a frequency of 2 MHz for signals of UMTS or WCDMA standard and a branch at 200 kHz for signals of GSM or DCS standard. Each filtering branch includes an input selection switch 6 associated with a filtering block 7. The selection of the active branch is performed by an appropriate control of the input selection switches 6, performed by the same control circuit as the selection switches 3 of input block 1. Finally, the reception chain includes a conversion block 8 (IF CONV) including as many coding branches, here, two, as block 6 includes filtering branches. Each branch of conversion block 8 includes an input switch 9 and an analog-to-digital converter 10. Converters 10 code over eight or twelve bits according to the standard used, respectively, GSM/DCS or UMTS/WCDMA. The selection of the active branch of conversion block 8 is performed in the same way as the selection of the active branch of filtering block 6.
Mixer block 4 should ideally be formed of a single multimode mixer. In practice, as will be detailed hereafter, several different mixing blocks 4 each corresponding to at least one mode to be processed must however be used. Thus, in the considered case of radiofrequency signals of standards GSM, DCS, UMTS, or WCDMA, at least two mixers are used. A first mixer processes the signals of mode GSM or DCS, and a second mixer processes the signals of standard UMTS or WCDMA. Generally, because of constraints of local oscillator LO, two separate mixers are used to process the signals of GSM mode and of DCS mode, which imposes the use of three separate mixers for the entire chain. To form each of the mixers, either a single-input mixer, associated with relatively complex common mode rejection circuits, or differential mixers, are used to automatically reject at least the common modes of even order.
FIG. 2A schematically illustrates the structure of a so-called Gilbert amplifier differential mixer. Such an amplifier includes, between two respectively high and low or reference voltage supply rails Vcc and GND, two input/output stages 20 and 30.
A first stage 20 (RF IN/OUT), hereafter the transconductance stage, receives the radiofrequency signal amplified by block 1 of FIG. 1. More specifically, input signal RF is provided to transconductance stage 20 in the form of two signals RF1 and RF2, one of which corresponds to the direct signal RF, the other one corresponding to signal RF shifted in phase by 180°. Each input signal RF1, RF2 is provided to a respective branch 21, 22 of transconductance stage 20. Each branch 21, 22 includes a respective NPN-type bipolar transistor 23, 24. The respective bases of transistors 23 and 24 form a first pair of input/output terminals of stage 20. Each input/output terminal receives, through a respective capacitor Cin1, Cin2, a single one of signals RF1 or RF2. The collectors of transistors 23 and 24 form a second pair of input/output terminals OUT1 and OUT2 of transconductance stage 20. The emitter of each transistor 23, 24 is connected, by a respective resistor R1, R2, to a first terminal of a source 25 of a D.C. current IDC (foot current) having its second terminal connected to reference power supply GND.
Second stage 30 (IF IN/OUT) includes a current-to-voltage conversion and filtering stage 31 and a switching stage 32. The second pair of input/output terminals OUT1/OUT2 of stage 20 also forms a pair of input/output terminals of switching stage 32. Terminal OUT1 is connected to a common emitter point of a first pair of NPN-type bipolar transistors 33 and 34. Terminal OUT2 is connected to the common emitter point of a second pair of identical NPN-type bipolar transistors 35 and 36. The bases of transistors 33 and 35 are interconnected and receive an oscillating signal LO1. The bases of transistors 34 and 36 are interconnected and receive another oscillating signal LO2. Signals LO1 and LO2 are at the frequency of local oscillator LO (FIG. 1), but phase-shifted by 180° with respect to each other.
The collectors of transistors 33 and 36 are interconnected. The collectors of transistors 34 and 35 are interconnected. The two obtained common points form a second pair of input/output terminals IF1, IF2 of second stage 30. The second pair of input/output terminals IF1, IF2 provides IF signal (FIG. 1) in the form of two signal phase-shifted by 180° with respect to each other. Each one of terminals IF1, IF2 is connected to high power supply Vcc through a respective branch of the current-to-voltage conversion and filtering stage 31 formed by the putting in parallel of a respective resistor R3, R4 and of a respective capacitor C3, C4. Resistors R3, R4 convert the output current of each branch into a voltage signal, of which respective capacitor C3, C4 filters the upper frequencies at the frequency of output signal IF1, IF2.
FIGS. 2B and 2C respectively illustrate characteristics of gain G and of current 1 versus the level of input voltage V in the mixer of FIG. 2A. The mixer gain is proportional to the product of gain Gm of transconductance stage 20 by input radiofrequency voltage VRF. The current of output signal IIF is equal to the sum of a static component equal to value IDC of D.C. current source 25, and of a dynamic component. The current of output signal IIF is limited by static component IDC. Under low supply voltages Vcc, the resulting mixer is then non-linear and output signal IF undergoes, with respect to input signal RF, a compression. This only slightly affects the processing of a signal of GSM or DCS standard in which, for a given power, the wanted data are coded in the signal by a phase modulation, the information about the signal amplitude being of no incidence. However, for signals of standard WCDMA, a given power envelope will include several messages corresponding to different power levels. The non-linearity of a Gilbert mixer then causes a loss of information.
To overcome this problem and to enable multimode operation, class AB mixers are used to process messages of standard WCDMA or UMTS.
The operating principles of a class AB mixer will be discussed hereafter in relation with FIG. 3A which illustrates the structure of a class AB amplifier and FIGS. 3B and 3C, homologous to FIGS. 2B and 2C, which illustrate the gain and output current characteristics according to the input voltage.
As illustrated in FIG. 3A, a class AB transconductance is essentially formed of a bipolar transistor 40 having its base forming an input/output terminal, receiving a radiofrequency signal IIN, having its collector forming an output terminal of a current IOUT, and having a degenerated emitter, that is, an emitter connected to reference power supply GND by an impedance 41. The radiofrequency signal is generally decoupled by a capacitor 42. Further, the base is connected to a bias source 43 by a resistor 44.
Such a transconductance exhibits an exponential characteristic of the gain according to the input base-emitter voltage, illustrated in FIG. 3B. This enables, as illustrated in FIG. 3C, obtaining an output current signal IOUT which varies exponentially as a function of the input signal, while a resistive base/emitter feedback causes no compression of the input signal, due to an increase in the static current. More specifically, the class AB transconductance exhibits a mean (static) current {overscore (IOUT)} which varies along with the input signal amplitude. If the base-emitter feedback is not very resistive, such a mean current causes no compression of the voltage difference Vbe between the base and the emitter. The dynamic component of output signal current IOUT is then no longer limited by bias signal IDC provided by source 25 but follows, or even exceeds, the mean current.
However, such a circuit samples, in the presence of a high input signal, a relatively high static current from the power supply. This is particularly disadvantageous in portable devices such as telephones or computers, since it requires frequent recharges of the device batteries. Further, such an assembly poses common mode rejection problems. The practical forming of the complete mixer, associating a first class AB non-differential transconductance stage of FIG. 3A with a second differential switching stage, is described for example in article “A class AB Monolithic Mixer for 900 MHz Applications” by K. L. Fong, C. D. Hull, and R. G. Meyer, published in IEEE Journal of Solid State Circuits, volume 32, No8, August 1997, which is incorporated herein by reference.